Switchable slot antenna

ABSTRACT

A compact, efficient and electronically tunable antenna is presented. A single-fed resonant slot loaded with a series of PIN diode switches constitute the fundamental structure of the antenna. The antenna tuning is real by changing its effective electrical length, which is controlled by the bias voltages of the solid state shunt switches along the slot antenna. Although the design is based on a resonant configuration, an effective bandwidth of 1.7:1 is obtained through this tuning without requiring a reconfigurable matching network. Four resonant frequencies from 540 to 890 MHz are selected in this bandwidth and very good matching is achieved for all resonant frequencies. Theoretical and experimental behavior of the antenna parameters is presented and it is demonstrated that the radiation pattern, efficiency and polarization state of the antenna remain essentially unaffected by the frequency tuning.

FIELD OF THE INVENTION

The present invention relates to a reconfigurable slot antenna having aplurality of shunt switches for changing an electrical length of theslot.

BACKGROUND OF THE INVENTION

With ever-increasing demand for reliable wireless communications, theneed for efficient use of electromagnetic spectrum is on the rise. Inmodern wireless systems spread spectrum signals are used to suppress theharmful effects of the interference from other users who share the samechannel (bandwidth) in a multiple-access communication system and theself-interference due to multipath propagation. Also spread spectrumsignals are used for securing the message in the presence of unintendedlisteners and alleviating the effects of communication jammers. Onecommon feature of spread spectrum signals is the relatively highbandwidth. This is specifically true for frequency-hopped spreadspectrum communications system. In a frequency-hopped spread spectrumsystem a relatively large number of contiguous frequency slots spreadover a relatively wide bandwidth are used to transmit intervals of theinformation signal. The selection of the frequency slots for each signalinterval is according to a pseudo-random pattern known to the receiver.

Signal propagation over large distances and in urban and forestedenvironment can take place at UHF and lower frequencies. At thesefrequencies, the size of broadband and efficient antennas isconsiderable. Techniques used to make the antenna size small, usuallyrenders narrow-band antennas. To make miniature size antennas compatiblefor a frequency-hopped spread spectrum system, we may consider areconfigurable narrow-band antenna that follows the pseudo-randompattern of the frequency-hopped modulation. The design aspects ofcompact, planar, and reconfigurable antennas are considered and thefeasibility of such designs is demonstrated by constructing and testinga planar reconfigurable slot antenna operating at UHF.

Compared to broadband antennas, reconfigurable antennas offer thefollowing advantages: 1) compact size, 2) similar radiation pattern andgain for all designed frequency bands, and 3) frequency selectivityuseful for reducing the adverse effects of co-site interference andjamming.

In recent years, reconfigurable antennas have received significantattention for applications in communications, electronic surveillanceand countermeasures by adapting properties to achieve selectivity infrequency, bandwidth, polarization and gain. In particular, preliminarystudies have been carried out to demonstrate electronic tunability fordifferent antenna structures. It has been shown that the operatingfrequency or bandwidth of resonant antennas can be varied when a tuningmechanism is introduced. Several interesting approaches are presented.In the literature, tuning is accomplished using varactor diodes, or bythe application of electrically and magnetically tunable substrates,with the use of barium strontium titanate (BST) and ferrite materialsrespectively.

Tuning of printed dipole or slot antennas has also been considered sincethey share the same advantages of portability, low profile andcompatibility in integration with other monolithic microwave integratedcircuits (MMICs). It has been shown in the literature that a 1 λ slotantenna can be tuned if loaded with reactive FET components. Althoughthe radiation pattern properties could be preserved in all resonantfrequencies, the tuning range of the resulting antenna was very limited.Second-resonance cross slot antennas have also been presented in amixer/phase detector system. In this application, a varactor diode wasused in the microstrip feed-line and the resonance could beelectronically tuned over a 10% bandwidth. This bandwidth was increasedto 45% when mechanical tuning was used by varying the feed-line length.Printed dipole tunable antennas have also been demonstrated loaded inseries with PIN diodes. The dipole length was varied from λ/2 to 1λdepending on whether the diodes were off or on. The operatingfrequencies were selected from 5.2 to 5.8 GHz, and only a very limitedmatching of 4-5 dB was achieved.

SUMMARY OF THE INVENTION

The slot antenna proposed in the present invention uses shunt switchesthat effectively change its electrical length over a very widebandwidth. To demonstrate the technique a reconfigurable slot antennacapable of operating at four different resonant frequencies over abandwidth of 1.7:1 is designed and tested. Measurements of the returnloss indicate that excellent impedance match can be obtained for allselected resonant frequencies. No special matching network is used andthe matching properties are solely determined by the placement of theswitches. The loading effect of the PIN diodes in the antenna is alsocharacterized by a full wave analysis and transmission line theory andcomparisons between the real and ideal switches are also studied. Perdesign goals, it is demonstrated that the reconfigurable slot antennahas the same radiation pattern at all frequencies. Also, the measuredradiation patterns agree with the theoretical ones. The polarizationcharacteristics and the efficiency behavior of the antenna as a functionof frequency are investigated using both theoretical and experimentaldata Finally, some design guidelines are provided and possible designimprovements are discussed.

The strict requirements of a constant input impedance, gain, radiationpattern and polarization can only be met, if both the passive structureand the tuning mechanism are carefully designed and effectivelyintegrated into the final design. Therefore, these issues are discussedseparately. First, the passive antenna structure and its properties arediscussed. The switching mechanism, its loading effect on the antennaand the final reconfigurable antenna are discussed next. Finally, themeasured results are presented.

Other applications of the present invention will become apparent tothose skilled in the art when the following description of the best modecontemplated for practicing the invention is read in conjunction withthe accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

The description herein makes reference to the accompanying drawingswherein like reference numerals refer to like parts throughout theseveral views, and wherein:

FIG. 1 is a resonant length at 600 MHz for straight slot antenna (infree-space wavelength) as a function of substrate thickness anddielectric constant;

FIG. 2A is a computed magnetic current distribution on 600 MHz straightslot antenna;

FIG. 2B is a computed magnetic current distribution on 600 MHz S-shapeslot antenna;

FIG. 2C is a computed magnetic current distribution on 700 MHz S-shapeslot antenna;

FIG. 2D is a computed magnetic current distribution on 600 MHz S-shapeslot with a short-circuit 21 mm above its bottom edge;

FIG. 3A is a S-shape slot antenna with microstrip feed-line;

FIG. 3B is the real and imaginary parts of the input impedance as afunction of frequency;

FIG. 4 is simulated results for the return loss of the S-shape slotantennas presented in FIGS. 2A through 2D;

FIG. 5A is a PIN diode connected as a shunt switch in a transmissionline;

FIG. 5B is a RF equivalent circuit for PIN diode including packagingeffects;

FIG. 5C is the isolation from the shunt diode used as a switch placed ina 60 Ω transmission line;

FIG. 6A is a layout of switch biasing network;

FIG. 6B is a RF equivalent circuit;

FIG. 6C is the On and Off-state simulated RF performance;

FIG. 7A is the RF equivalent circuits for determining the resonantfrequency of an unloaded single switch slot antenna;

FIG. 7B is the RF equivalent circuits for determining the resonantfrequency of a loaded single switch slot antenna;

FIG. 8A is a slot antenna with resistive load representing actuatedswitch (units are in mm);

FIG. 8B is the return loss for different values of switch resistance;

FIG. 8C is the improved return loss withminor adjustments (<4 mm) in theslot length above the feeding point.

FIG. 9A is the reconfigurable slot antenna (units are in mm);

FIG. 9B is the simulated return loss for the four resonant frequencies;

FIG. 9C is the typical radiation pattern;

FIG. 9D is the simulated gain the four resonant frequencies;

FIG. 10 is the measured resonant frequencies of the reconfigurableantenna; and

FIGS. 11A through 11D are the measured radiation patterns for the fourresonant frequencies.

DESCRIPTION OF THE PREFERRED EMBODIMENT

The antenna size at UHF and lower becomes critical and therefore specialconsideration is required. A compact planar geometry is best suitedsince three-dimensional large and bulky structures are in generalundesirable. Furthermore, some miniaturization techniques have beenapplied to reduce the size. This section focuses on the passive slotantenna design issues emanating from the above principles.

First, the miniaturization capabilities provided by a high dielectricconstant substrate were investigated. Inasmuch as an accuratecharacterization of its effect is needed, a commercially availablemoment method code was employed. First, simple slot antennas weresimulated at 600 MHz and their resonant lengths were determined as afunction of the substrate thickness and dielectric constant (FIG. 1).This analysis suggests that even at low frequencies where the substrateis very thin compared to the wavelength, a miniaturization factor ofabout 2:1 is possible, if a high dielectric constant substrate isemployed. However, the standard commercially available substrates areelectrically thin at UHF and below and therefore the 2:1 factor seems tobe a limit difficult to exceed even for substrate permittivities as highas 10.

In an effort to further decrease the total area occupied by the antenna,the slot configuration was altered from its standard straight form to anS-shape. From the simulated equivalent magnetic current distribution onthe straight and S-shape slots (FIGS. 2A and 2B), it is obvious that thedistributions both closely follow a sinusoidal pattern with the maximumcurrent concentrated in the middle of the slot. As a result, the twoantennas share very similar properties and only differ in thepolarization orientation. The antenna of FIG. 2A is horizontallypolarized, while the antenna of FIG. 2B slant linearly polarized. Othermore complicated geometrical shapes can also be used, but the S-shapeslot does not contain any segments supporting opposing currents, whichwould considerably deteriorate the radiation efficiency. It should alsobe mentioned that, although the total area of the antenna is greatlyreduced by this geometrical change, the resonant length remains almostunchanged. For example, a resonant length of 136 mm for a straight slotis slightly increased to 139 mm for S-slot at 600 MHz for a substratewith e, =10.2 and thickness of 2.54 mm.

The standard microstrip feed for the simple slot can also be used forthe S-shape slot. FIG. 3A shows the slot antenna with its feed-line,while FIG. 3B presents the input impedance at the feeding point as afunction of frequency. To achieve a good match to a 50 Ω line, themicrostrip feed-line has to be moved close to one end of the slotantenna. This implies that the antenna input impedance is not verysensitive to small changes in the length of the longer segment (12, seeFIG. 3A). This property will greatly simplify the design of the tunableslot and its feeding network and will result in minimum complexity andmaximum reliability for the final antenna. This property of the slotantenna makes it an attractive choice as a reconfigurable structure,since most other antennas (such as dipoles) would require a speciallydesigned matching network.

The resonant frequency of the above structure can be tuned by changingthe electrical length of the slot. This may be readily accomplished byintroducing a short circuit at a specific location. Then the slot willappear to be shorter and therefore the antenna will resonate at a higherfrequency. The three S-shape slots in FIG. 2B through 2D demonstratethese concepts. The slot antenna of FIG. 2B resonates at 600 MHz with aresonant length of 139 mm. The antenna of FIG. 2C is 21 mm shorter andis designed to resonate at 700 MHz Finally, the antenna of FIG. 2D isobtained by modifying the antenna of FIG. 2B. In particular, the antennaof FIG. 2B is short circuited at 21 mm above its lower end. Thesimulated return losses for these three slots are shown in FIG. 4. It isalso important to note that the microstrip feed-line remains unchangedin all three cases. That is, the distance between the top end of theslot and the feed line cross point remains constant and is equal to 3.2mm. This means that, although the resonant frequency is shifted by 100MHz, very good matching is achieved for both slot antennas illustratedin FIGS. 2C and 2D without the need for modifying the feeding network.In addition, the slot antennas of FIGS. 2C and 2D have almost identicalresonant frequencies. The small difference in the resonant frequencycomes from the fact that the antenna of FIG. 2D appears somewhatelectrically longer than the antenna of FIG. 2C due to the parasiticeffects of the short circuit. Therefore, tunability is possible byintroducing these short circuits with no special matching network.Although FIG. 2A through 2D illustrates the basic concept ofreconfigurability on a dual band antenna, it is obvious that it can beextended to antennas with several bands of operation. The number ofthese bands depends on the number of switches on the antenna. Forexample, a four band antenna is presented and it is demonstrated thatthe resonant frequency can be digitally controlled by an array of fourswitches.

The basic principle of controlling the antenna resonant frequency hasbeen discussed above. It was also shown that even when a perfect shortcircuit is used, the parasitic effects of the short can slightly affectthe antenna performance and particularly the resonant frequency. Theparasitic effects become worse when a switch with finite isolation isused. The issues related to the design of a suitable solid state switchand on the characterization of its effects on the antenna performancewill now be discussed. Finally, the complete reconfigurable antennadesign is presented below together with its theoretical performance.

A. Switch Design

To implement the electronic reconfigurability, the ideal shunt switchesmust be replaced with PIN diodes. PIN diode's reliability, compact size,high switching speed, small resistance and capacitance in the on and offstate respectively make it most appropriate for the application at hand.The RF equivalent circuit of the diode is shown in FIG. 5B for both theon and off states. The reactive components C_(p) and L_(p) model thepackaging effect, while the others come from the electric properties ofthe diode junction in the on and off positions. Typical values are alsogiven for the HSMP-3860 diode used in the present invention. Thecomputed isolation (defined as 1|S₂₁|²) for the circuit shown in FIG. 5Ais given by: $\begin{matrix}{\alpha = {10\quad{\log\lbrack \frac{( {\frac{R_{d}Z_{0}}{R_{d}^{2} + X_{d}^{2}} + 2} )^{2} + ( \frac{X_{d}Z_{0}}{R_{d}^{2} + X_{d}^{2}} )^{2}}{4} \rbrack}}} & (1)\end{matrix}$where Z_(d)=R_(d)+jX_(d) is the equivalent impedance of the diode and Z₀is the characteristic impedance of the line. In the example consideredhere, the characteristic impedance of the line is approximately equal to60 Ω, which is calculated by the moment method code for a slotline witha width of 2 mm, a finite ground plane of 60 mm (on both sides of theslot) and a substrate permittivity ε_(r)=10.2 (RT/Duroid). The isolationcomputed in equation (1) is plotted in FIG. 5C as a function offrequency for the HSMP-3860 diode in the 60 Ω slotline. Althoughisolation greater than 25 dB is possible at low frequencies, it degradesto 17 dB at 600 MHz and only 11 dB at 1 GHz due to the diode parasiticelements. However, as will be shown, this attenuation is sufficient fora successful antenna tuning up to 900 MHz.

The switch bias network is presented in FIGS. 6A through 6C. An inductorof 470 nH and three 10 pF capacitors are used to improve the RF-DCsignal isolation. These values were chosen based on the bias network RFequivalent circuit shown in FIG. 6B. The simulated performance for theon and off states is presented in FIG. 6C. The RF-DC isolation is betterthan 30 dB for both states and the return loss is less than −20 dB forthe off state. Finally, the RF-RF isolation is comparable to the oneshown in FIG. 5C.

B. Switch Loading on the Antenna

Although the switch isolation is important since it determines thefrequency selectivity of the antenna, the switch loading on the antennais equally important inasmuch as it affects its resonant frequency andinput impedance. The loading effects must be taken into account for anaccurate prediction of the antenna resonant frequency and inputimpedance, especially when more than one switch is used formulti-frequency operation.

A transmission line equivalent circuit that models the loading effect ofone diode on the antenna is shown in FIGS. 7A and 7B. The transverseresonant technique states that:Z _(R)(z′)+Z _(L)(z′)=0  (2)where Z_(R)(z′) and Z_(L)(z′) are the input impedances on the right andleft of the reference point respectively. For the unloaded transmissionline in FIG. 7A equation (2) simplifies to: $\begin{matrix}{{{{\tan( {\beta\quad l_{L}} )} + {\tan( {\beta\quad l_{R}} )}} = 0}{or}} & (3) \\{{{\beta( {l_{L} + l_{R}} )} = {n\frac{\pi}{2}}},{n = 1},2,{3\quad\cdots}} & (4)\end{matrix}$which is the well known formula for these resonant antennas. Now it isimportant to see what happens in the simplest case of having one switchon the antenna FIG. 7B shows the equivalent circuit of a transmissionline loaded with one switch in the off position. Equation (2) thenbecomes:[Z ₀ω_(R) C−cot(βl _(R2))][tan(βl _(L))+tan(βl _(R1))]=1+tan(βl_(L))tan(βl _(R1))  (5)Equation (5) can of course be solved numerically and an iterative methodcan be employed for finding the unknown lengths until the desiredresonant frequency (f_(R)) has been achieved. A similar procedure can befollowed if more than one switch is used on the slot, but the processbecomes a little more complicated if all resonant frequencies are to bespecified. We also need to note that equation (5) does not include anypackaging effects, but these can be readily incorporated in the model,resulting in a more accurate computation.

Only the loading effects when the switch is in its off state have beendiscussed up to now. Nevertheless, the small on-state resistance alsoaffects the antenna performance and particularly its input impedance.Full wave analysis was used to model these effects. For a first orderapproximation, the resistance was modeled as a thin film resistor on topof the slot and the packaging parasitic elements were neglected in thisanalysis. The parasitic element effects in the on-state can be importantespecially at the highest frequencies (see FIG. 5C). FIG. 8A shows thesimulated geometry of an S-shape slot antenna loaded with a resistivefilm, which is fed by a microstrip line and FIG. 8B shows the simulatedreturn loss versus the switch on-state resistance for four differentcases between 0 to 5.6 Ω. In all four cases the position of the 50 Ωfeed-line was kept unchanged. It is obvious that the matching leveldeteriorates rapidly as the resistance value increases, and forresistance values above 1.5 Ω the matching level becomes unacceptable.

However, this degradation can be avoided to some extent by elongatingthe upper end of the slot as the resistance is increased. FIG. 8C showsthe improvement on the antenna matching when the slot length isadjusted. It is found that, in all three cases, only a very small linesegment length needs to be added in order to improve the input impedanceof the antenna. Even for a resistance value of 5.6 Ω the required linesegment length is less than 3% of the total slot length, resulting inonly a small change in the resonant frequency. This method ofmaintaining a good impedance match will be used later for the design ofthe reconfigurable antenna by placing additional switches(matching-switches) on the slot above the feed-line and synchronizingthem together with the switches at the other end of the slot(frequency-switches). However, it should be noted that the matchingswitches will not represent perfect shorts and they will introduce anextra loading effect. Nonetheless, this effect is negligible andmatching levels of better than −20 dB can be achieved, as will be seennext. Therefore, the matching properties of the reconfigurable antennawill solely depend on the position of an array of switches on the slotand no matching network will be necessary as frequency changes. TABLE ICOMPUTED EFFICIENCY FOR SLOT ANTENNAS WITH A SINGLE SWITCH VERSUSON-STATE RESISTANCE VALUE R [Ω] 0 1.4 2.8 5.6 Efficiency [1%] 71.8 55.645.6 33.9

Having discussed the loading effects of the switches on the matchingproperties of the antenna, the effects on the radiation characteristicsof the antenna need to be found as well. Ideally, the radiationefficiency should be that of the half-wavelength dipole, since theantenna behaves effectively as a λ/2 resonant slot at each of itsoperating frequencies. However, the on-state resistance of the switcheswill obviously result in power dissipation and finally degradation inthe antenna efficiency. The dissipated power obviously depends on thediode's on-resistance and on the number of the switches on the antenna.Table I shows the computed efficiency for the antennas previouslydiscussed in FIGS. 8A through 8C. Dielectric loss has been included inall cases. This explains the non-ideal efficiency when R=0 Ω.

The above antenna efficiency analysis shows that even for a small seriesresistance of R=1.4 Ω the antenna gain will be approximately 2.5 dBlower than that of an ideal halfwavelength dipole. This is an inherentdrawback of using PIN diode switches. However, micro-electro-mechanical(MEMS) switches are becoming increasingly important and are now a viablealternative as the MEMS offer very low power consumption and the MEMScome even in smaller packages. It has also been shown that capacitivetype MEMS switches exhibit very low ohmic losses and therefore can beused for maximized antenna efficiency. However, the requiredon-capacitance values renders the capacitive type MEMS impractical forUHF frequencies. Hence metal-to-metal contact switches, which have nocut-off frequency should be considered in such a design.

C. Final Reconfigurable Antenna Properties

Based on the previously discussed design principles, a reconfigurableslot antenna design (shown in FIG. 9A) is presented here. Four switchesare used in order to tune the antenna over a range of 540 to 950 MHz.Both full wave analysis and the transmission line model TABLE IITHEORETICALLY CALCULATED RESONANT FREQUENCIES USING FULL WAVE ANALYSISAND TRANSMISSION LINE MODEL f_(R) [MHz] f_(R) [MHz] (TLN)^(a) (MM)^(b)Switch Configuration 542 561 4 = ON 1, 2, 3, = OFF 596 627 1, 4 = ON 2,3 = OFF 688 711 2, 4 = ON 1, 3 = OFF 1002 950 3 = ON 1, 2, 4 = OFF^(a)Transmission Line Model^(b)Moment Methodwere used in the design process. In this design three frequency-switchesand a single matching-switch are used. Table II summarizes thecalculated resonant frequencies and the conditions of all four switchesfor each resonant frequency. The transmission line model has theadvantage of allowing fast and accurate (as will be proven later)computation of the resonant frequencies and can easily incorporate thediode parasitics. However, the full wave analysis is essential when anaccurate prediction of the antenna input impedance is needed. In themoment method code, the diodes were simulated as metal-insulator-metal(MIM) capacitors and as thin film resistors in the off and on statesrespectively and as a result the packaging parasitics were ignored. Thisexplains the 5% differences observed in the computed resonances betweenthe two models. FIG. 9B shows the calculated return loss where amatching level of better than −20 dB has been achieved for all theoperating frequencies.

Since at every operating frequency the antenna radiates as a λ/2 slot,the radiation pattern remains unchanged when the frequency is shifted.The same holds for the antenna directivity. The E and H-planes of atypical calculated pattern are shown in FIG. 9C. Since the antenna hasbeen designed on a electrically thin substrate (at UHF) the radiationpattern is symmetric on the two sides of the slot However, theefficiency and the gain will be reduced compared to a half-wavelengthdipole due to the resistive losses caused by the TABLE III CALCULATEDPOLARIZATION FOR THE RECONFIGURABLE ANTENNA f_(R) [MHz] 561 627 711 950Angle (°) 60 70 60 40diodes. FIG. 9D shows the calculated gain using the moment methodanalysis. The gain is approximately −1 dB for the lowest frequencies andincreases to about 0.7 dB for the highest one. Similar results hold forthe antenna efficiency.

The reference angle of 0° in the previous graphs represents thedirection normal to the antenna ground plane. Although the S-shapepattern considerably reduces the antenna occupied area, it has theinherent drawback that the polarization does not remain constant as thefrequency is changed. However, as Table III shows, the polarization doesnot change considerably (variation of about 30°). This is due to thefact that the antenna polarization (always slant linear) is dominated bythe orientation of the middle segment of the slot where most radiatedfield is emanated from. Therefore, if the orientation of the receivingantenna does not follow that of the transmitter as the frequency ischanged, a maximum polarization mismatch of 25% will be incurred. Theorientation of linear polarization reported in Table III is with respectto the x-axis (see FIG. 9A).

The reconfigurable antenna designed in the previous section wasfabricated on a 100 mil thick RT/Duroid substrate (ε_(r)=10.2). The sizeof the ground plane was 5×5 in².

The first task was to measure the resonances and an HP8753D vectornetwork analyzer was used for the S-parameter measurements. The biasingvoltage for the switches was provided by a DC voltage source. Aftercalibrating the network analyzer the antenna return loss was measuredwhen different Combinations of the switches were activated. The measureddata are presented in FIG. 10, where a return loss of better than −13 dBis observed at all resonances. The measured resonances are shown inTable IV together with the necessary biasing conditions. Satisfactoryagreement between theoretical, Table II, and experimental, Table IV,data is observed. In addition, the transmission line model TABLE IVMEASURED RESONANT FREQUENCIES AND THE NECESSARY BIAS VOLTAGES FOR THESWITCHES Bias Voltage [V] f_(R) [MHz] S1 S2 S3 S4 537 −20 −20 −20 1.1603 1.1 −20 −20 1.1 684 0 1.1 −20 1.1 887 0 1.1 1.1 0.2gives slightly better results—except the highest frequency—mainlybecause the parasitic reactive elements have been included in this modeland not in the moment method technique. However, this is not true forthe highest resonant frequency where an error of 13% exist between thetransmission line model and the measurement. This discrepancy can beattributed to the fact that the properties of the diodes, andparticularly the element values of its equivalent circuit, cannot beassumed constant up to 1 GHz.

A reverse voltage of −20 V was applied to maintain the switches in theoff position and by doing so a better matching level was achieved. Thisis an important issue particularly when the antenna is used as thetransmitter. Since the structure is a resonant structure strong electricfields are established that can turn the diodes on and off at the RFfrequency and ruin the small signal design. This effect was clearlyobserved at the lowest resonance with an input power of 0 dBm. In thiscase an improvement of about 5 dB was achieved by changing 0 V bias to−20 V.

One more interesting effect was observed for the highest resonance.Better matching level would occur, if not only S3 but also S2 wasforward biased. This is due to the relatively low isolation that eachdiode provides at these relatively high frequencies (see FIG. 5C).Therefore, biasing S2 results in higher isolation and reduces the effectof leaked magnetic current in the area after the switch The improvementin the return loss was approximately 10 dB compared to leaving S2unbiased for this frequency.

Next, far field patterns were measured in the University of Michigan'sanechoic chamber. The E and H-plane were measured as well as thecorresponding cross-polarization for each operating frequency. An RFsignal and a DC voltage source were used with the reconfigurable antennaand a dipole with adjustable length was employed as the receivingantenna. The dipole length was appropriately adjusted for each operatingfrequency of the transmitting antenna until maximum received power wasrecorded. In order to find the Eplane, the transmitting antenna wasrotated until the electric field was vertically polaried. Then thetransmitter, placed on a turn table, was azimuthally rotated formeasuring Eplane cuts. The cross-polarized pattern was measured byrotating the receiver antenna by 90°. H-plane pattern measurements wereconducted in a similar manner.

Although for slot antennas printed on a substrate it is expected thatthe radiated power be higher in the half-space that include thedielectric substrate, no appreciable difference was observedexperimentally. This is easily explained since in this case thedielectric thickness is about λ/200 at 600 MHz and the size of theground plane is small (approximately λ/3) at the same frequency.Therefore the antenna is almost bi directional and equivalent to adipole in free space.

The measured data are presented in FIGS. 11A through 11D for eachresonant frequency. In these plots 0° denote the direction of maximumradiated power. These measurements show that the H-plane closely followsthe expected sinusoidal pattern. However, some slight asymmetries near±90° exist for almost all frequencies. These discrepancies originateprimarily from two sources. First, parasitic radiation from the cablesand the feeding network and second, radiation from the edges of thedielectric. These sources of radiation also affected the Eplane patternmeasurements and caused a difference of 3-4 dB between the minimum andmaximum measured value. (see FIGS. 11A through 11D). Despite thesediscrepancies, it is clear that the far-field pattern remains unchangedversus the frequency tuning.

Gain measurements are accomplished using the comparison method. Alog-periodic antenna with 6 dBi gain at 600 MHz was used as a referenceantenna for these measurements. The second resonance at 593 MHz waschosen as the operating frequency of the reconfigurable antenna, so thatto make direct comparisons with the reference antenna possible. Tomeasure the gain, the power received by the receiver dipole at 593 MHzwas recorded when both the reference and the reconfigurable antennaswere used in the TABLE V MEASURED POLARIZATION FOR THE RECONFIGURABLEANTENNA f_(R) [MHz] 537 603 684 887 Angle (°) 57 70 55 33transmitting mode inside the anechoic chamber under the same conditions.The measured gain was found −1.1 dBi, which corresponds to an efficiencyof 47%. These results closely resemble the calculated data It shouldalso be pointed out that the gain of the slot antenna is reduced notonly from the forward-biased diode resistance, but also from the smallground plane size. However, a comparison between the measured andcalculated data reveals that the dominant degrading factor in gain isthe dissipated power on the diodes rather the ground plane size.

Finally, the antenna polarization was measured and the method previouslydescribed for the pattern measurement was employed. The measuredpolarization orientation at each frequency is provided in Table V. Asdiscussed before, although the polarization does not remain absolutelyconstant as the frequency is changed, the variation range is small andcomparable to the theoretical data (see Table III).

A novel method for designing affordable, compact, reconfigurableantennas is proposed in the present invention. This method relies onchanging the effective length of a resonant slot antenna by controllingcombinations of electronic RF switches. Theoretical results forsignificant antenna parameters were validated experimentally. Importantissues involved in the design of such antennas and guidelines were alsodiscussed. Based on the proposed method, a compact planar reconfigurableslot antenna was designed, fabricated and measured and a tuning range of1.7:1 in the operating frequency was demonstrated. Although such a broadrange was achieved, no matching network was required for the antenna.Another salient feature of this design, backed by theory andexperiments, is that the radiation characteristics of this antennaremain essentially unaffected by the frequency tuning. The designprocedure is general enough and allows even wider tuning ranges to beachieved. By employing suitable switches it can be also readily extendedto higher frequency applications.

While the invention has been described in connection with what ispresently considered to be the most practical and preferred embodiment,it is to be understood that the invention is not to be limited to thedisclosed embodiments but, on the contrary, is intended to cover variousmodifications and equivalent arrangements included within the spirit andscope of the appended claims, which scope is to be accorded the broadestinterpretation so as to encompass all such modifications and equivalentstructures as is permitted under the law.

1. A slot antenna comprising: a substrate having a single fed resonantslot formed therein; and a plurality of shunt switches for electricallychanging an electrical length of the slot over a wide bandwidth.
 2. Theslot antenna of claim 1 further comprising: the slot antenna operatingat a plurality of different frequencies over a bandwidth ofapproximately 1.7:1.
 3. The slot antenna of claim 1 wherein the shuntswitches further comprises: a series of PIN diode switches loaded on theslot.
 4. The slot antenna of claim 1 wherein the shunt switches furthercomprises: a series of micro-electro-mechanical switches loaded on theslot.
 5. The slot antenna of claim 1 wherein the shunt switches furthercomprises: at least one matching switch loaded on the slot.
 6. The slotantenna of claim 1 wherein the shunt switches further comprises: atleast two frequency switches loaded on the slot.
 7. The slot antenna ofclaim 1 wherein the shunt switches further comprise: at least one PINdiode switch in an off position and being subjected to a reverse voltageto maintain the switch in the off position.
 8. The slot antenna of claim1 further comprising: the slot formed in the substrate having an S-shapedefined by three portions, where two outer portions are angled atapproximately 90° with respect to one transversely extending innerportion connecting opposing ends of the outer portions.
 9. The slotantenna of claim 8 further comprising: the plurality of shunt switchesforward biasing two of the three portions of the S-shape slot.
 10. Theslot antenna of claim 9 further comprising: the two forward biasedportions include one of the outer portions and the inner portion of theslot.
 11. The slot antenna of claim 1 further comprising: means foroperating at different polarizations for different frequencies within anoperating bandwidth of the slot antenna.
 12. The slot antenna of claim 1further comprising: a very high selectivity over an entire operatingbandwidth of the slot antenna allowing operation in reconfigurablewireless networks and anti-jamming systems.
 13. The slot antenna ofclaim 1 further comprising: means for obtaining a different polarizationfor every frequency in an operating bandwidth of the slot antenna byappropriately positioning switches on each segment of the slot.
 14. Aslot antenna comprising: a substrate having a single fed resonant slotformed therein; and means for changing an effective length of theresonant slot by controlling combinations of electronic radio frequencyswitches.
 15. A method for designing a slot antenna comprising the stepsof: providing a substrate having a single fed resonant slot formedtherein; and changing an effective length of the resonant slot bycontrolling combinations of electronic radio frequency switches.
 16. Themethod of claim 15 further comprising the step of: electrically changingthe effective length of the slot over a wide bandwidth with a pluralityof shunt switches.
 17. The method of claim 15 further comprising thestep of: operating the slot antenna at a plurality of differentfrequencies over a bandwidth of approximately 1.7:1.
 18. The method ofclaim 15 further comprising the step of: loading a series of PIN diodeswitches on the slot.
 19. The method of claim 15 further comprising thestep of: loading a series of micro-electro-mechanical switches on theslot.
 20. The method of claim 15 further comprising the step of: loadingat least one matching switch on the slot.
 21. The method of claim 15further comprising the step of: loading at least two frequency switcheson the slot.
 22. The method of claim 15 further comprising the step of:subjecting at least one PIN diode switch in an off position to a reversevoltage to maintain the switch in the off position.
 23. The method ofclaim 15 further comprising the step of: forming the slot in thesubstrate with an S-shape defined by three portions, where two outerportions are angled at approximately 90° with respect to onetransversely extending inner portion connecting opposing ends of theouter portions.
 24. The method of claim 23 further comprising the stepof: forward biasing two of the three portions of the S-shape slot. 25.The method of claim 24 further comprising the step of: selecting the twoforward biased portions to include one of the outer portions and theinner portion of the slot.
 26. The method of claim 15 further comprisingthe step of: operating at different polarizations for differentfrequencies within an operating bandwidth of the slot antenna.
 27. Themethod of claim 15 further comprising the step of: providing very highselectivity over an entire operating bandwidth of the slot antennaallowing operation in reconfigurable wireless networks and anti-jammingsystems.
 28. The method of claim 15 further comprising the step of:obtain a different polarization for every frequency in an operatingbandwidth of the slot antenna by appropriately positioning switches oneach segment of the slot.